circuit harness wiring

For a collector follower with emitter resistor, you’ll often find that the gain per stage is no more than 10 to 50 times. The gain increases when the emitter resistor is omitted. Unfortunately, the distortion also increases. With a ubiquitous transistor such as the BC547B, the gain of the transistor is roughly equal to 40 times the collector current (Ic), provided the collector current is less than a few milliamps. This value is in theory equal to the expression q/KT, where q is the charge of the electron, K is Boltzmann’s constant and T is the temperature in Kelvin.

For simplicity, and assuming room temperature, we round this value to 40. For a single stage amplifier circuit with grounded emitter it holds that the gain Uout /Uin (for AC voltage) is in theory equal to SRc. As we observed before, the slope S is about 40Ic. From this follows that the gain is approximately equal to 40I cRc. What does this mean? In the first instance this leads to a very practical rule of thumb: that gain of a grounded emitter circuit amounts to 40·I c·Rc, which is equal to 40 times the voltage across the collector resistor.

If Ub is, for example, equal to 12 V and the collector is set to 5V, then we know, irrespective of the values of the resistors that the gain will be about 40R(12–5) = 280. Notable is the fact that in this way the gain can be very high in theory, by selecting a high power supply voltage. Such a voltage could be obtained from an isolating transformer from the mains. An isolating transformer can be made by connecting the secondaries of two transformers together, which results in a galvanically isolated mains voltage.

That means, that with a mains voltage of 240 Veff there will be about 340 V DC after rectification and filtering. If in the amplifier circuit the power supply voltage is now 340 V and the collector voltage is 2 V, then the gain is in theory equal to 40 x (340–2). This is more than 13,500 times! However, there are a few drawbacks in practice. This is related to the output characteristic of the transistor. In practice, it turns out that the transistor does actually have an output resistor between collector and emitter.

This output resistance exists as a transistor parameter and is called ‘hoe’. In normal designs this parameter is of no consequence because it has no noticeable effect if the collector resistor is not large. When powering the amplifier from 340 V and setting the collector current to 1 mA, the collector resistor will have a value of 338 k. Whether the ‘hoe’-parameter has any influence depends in the type of transistor. We also note that with such high gains, the base-collector capacitance in particular will start to play a role.

As a consequence the input frequency may not be too high. For a higher bandwidth we will have to use a transistor with small Cbc, such as a BF494 or perhaps even an SHF transistor such as a BFR91A. We will have to adjust the value of the base resistor to the new hfe. The author has carried out measurements with a BC547B at a power supply voltage of 30 V. A value of 2 V was chosen for the collector voltage. Measurements confirm the rule of thumb. The gain was more than 1,000 times and the effects of ‘hoe’ and the base-collector capacitance were not noticeable because of the now much smaller collector resistor.

Author: Gert Baars Copyright: Elektor Electronics

What’s more, if unit growth continues for photovoltaic and other energy-harvesting systems relying on large-capacity storage batteries, designs that fail to maximize battery life could have a negative environmental impact due to the extra material and energy consumption needed to manufacture replacement systems as well as dispose of exhausted units.

The latest addition to my collection of Infra Red (IR) Repeater circuits. The Mark 5 is a much improved version of the Mark 1 circuit and has increased range and sensitivity. It is also immune to the effects ofambient light, daylight and other forms of interference. In addition it works with IR modulation freuencies in the range 30 to 120kHz making the Mk5 circuit the best choice for compatibility with remote controls.

Parts List:
R1,R2: 5M6 RESISTOR (2)
R3,R5: 3k3 RESISTOR (2)
R4: 120k RESISTOR (1)
R6: 220R RESISTOR (1)
R7: 47k RESISTOR (1)
R8: 120R RESISTOR (1)
R9: 10k RESISTOR (1)
R10: 2K2 RESISTOR (1)
R11: 100R RESISTOR 1 W (1)
C1,C3,C4: 22n polyester CAP (3)
C2: 100u electrolytic 25V(1)
C5: 100u electrolytic 25V(1)
Q1 BC107 (1) alternatives, BC107A, 2N2222, 2N2222A
Q2 BC109C (1) alternatives, BC109, BC549
D1: 1N4148 DIODE (1)
D2: Red LED (1)
IC1,IC2 CA3140E opamp (1)
IR1: SFH2030: (1)
IR2,3: TIL38 (2) or similar.

Design Philosophy:
This time I have returned to "first principles" and built a wideband infra red (IR) preamp which receives and re-transmits the entire baseband signal from a remote control handset.

It is designed to work with IR controls using 30-120KHz and should therefore work with just about any handset. In addition I have separated ambient (surrounding) light from the modulated light used by a remote handset. The major problem with the Mark 1 circuit is that it reacts to all light sources, ambient light producing a continous signal from the IR photo diode and is amplified by the rest of the circuit. I have published a modification to the original Mark 1 circuit, click here to view.

Noise Immunity:
It is difficult working with Infra Red, you cannot see it, and it is difficult to measure. A major barrier with this circuit was how to differentiate between daylight and an IR signal. Ambient light produces an almost continuous signal, changing little over several hours. A signal from an IR handset contains control pulses modulated with a carrier frequency (typically 36kHz) transmitted using an Infra Red photo diode. My solution used here, is a simple RC filter formed by C1 and R3.

At low frequency i.e. 50Hz the impedance of C1 is high, around 144k. The voltage gain of inverting op-amp IC2 is approximately R4 / R3, but at low frequency C1 is in series with R3 so the gain is now 120k / (3.3k + 144k) or less than unity. Daylight or ambient light will change slowly over several hours, in frequency terms this signal would be millihertz or less and C1s impedance will be megaohms.

A signal from an IR handset will be modulated at around 36KHz. At this frequency the impedance of C1 is very low, around 200 ohms. This has little effect on the input impedance of the op-amp stage and voltage gain will now be R4 / R3 or about 34 times. The impedance of capacitor C4 also helps noise rejection as its impedance change will allow more signal to pass into Q1 base at high frequencies and much less signal at line frequencies.

Circuit Details:
Light photons are received at IR1, this is an IR photo diode type SFH2030. A SFH2030F, which contains a daylight filter,may also be used instead of the SFH2030. The photo diode is reverse biased and when light strikes it, the energy of the IR signal releases additional charge carriers within the diode, allowing more current to flow. This current is amplified and converted to a voltage by the first CA3140 opamp, IC1. IC1 is wired as a current to voltage convertor, see below.

In an ideal current to voltage convertor the output voltage would be the product Rf multiplied by the input current. The non-inverting input would be tied to ground. In the Mark 5 circuit the output voltage is iR1 or about 5.6 Volts/uA appearing at pin 6 of IC1. The current generated by the SFH2030 photo diode when receiving a signal from a handset several metres away is less than 50 nA and requires the extreme high input impedance to avoid shunting the signal. There are two reasons for using the CA3140, the first is its high input impedance, over 1000G. The second reason is that normally the non-inverting input would be at 0V when working from split + and - supplies. In this single supply version the non-inverting input is returned to negative supply via R2. This can only be done with a Mosfet input, hence the choice for using the CA3140.

IC1 converta all current from the photo diode IR1 into a voltage. Although the SFH2030 is most sensitive at infra red wavelengths, it will produce tiny currents from daylight and also the 50/60Hz noise fields from flourescent and mains lighting. To minimize this, C1 and R3 form a high pass filter, allowing a 30kHz and higher signals to pass but blocking low frequencies. The impedance of C1 increases with decreasing frequency being 31k at 50Hz. Daylight for example, produces a contstant luminence, changing slowly over several hours, to which the impedance of C1 is effectively infinite.

The signal voltage from IC1 is now further amplified by IC2, gain being the ratio R4/R3 or 31dB. All opamps have a limit called the gain bandwidth product. The gain will fall to unity at the highest usuable frequency and be a maximum value at dc. Between these limits the gain falls with increasing frequency as shown in the bode plot for the CA3140 below:


Looking at the chart above, at 100kHz the maximum gain can only be about 30dB. However this is ample and boosts the received range of signals from a remote handset to the photo diode which have worked well up to 4 metres apart. Because R5 is returned to the negative supply a Mosfet input opamp must again be used. The output is again filtered by a high pass filter comprising C4 and the associated input impedance of Q1. R6, C2 and C3 provide decoupling for the IR preamplifier, C3 is in parallel with C2 because an electrolytic is not always a low impedance at high frequencies.

The IR output stage is comprised of Q1 and Q2 and associated components. The output is arranged so that with no input signal, Q1 is on and Q2 off; the visible LED, D2 will also be off. With no signal the 47k resistor biases the driver transistor, Q1 into full conduction. Its collector voltage will be near zero volts and the output transistor Q2, which is direct coupled to Q1 collector will therefore be fully off. Power drain will be minimal.

When an IR signal is receieved from a handset, the complete modulated signal will be amplified and fed via C4 into Q1 base. This is sufficiently strong enough to overcome the positive bias supplied by R7 and switch off Q1. This will happen many times a second, at the same frequency as the IR modulating signal sent by the handset. As Q1 switches off, its collector voltage rises to near full supply switching on Q1 and lighting the LED D2. Pulses of infra red at the same modulating frequency are then transmitted by the photo emitting diodes, IR2 and IR3. Because the signal is cleaner, (i.e. no daylight or 50/60Hz lamp fields included) then the series resistor R11 has been incresed in value to 100 ohms. The range from photo emitter diode to the equipment to be controlled has proved successfull at over 4 metres when powered from a 12 Volt supply. D1 helps to improve the turn off speed of Q1, thereby ensuring that the output waveform will be "squarer". It can be omitted but the circuit will perform better if D1 is included. A simulated transfer characteristic is shown below:

AC Transfer Charcteristic


The ouput is measured between Q2 emitter and ground. A simulated transient response is shown below. Three graphs are produced with excitations of 40,80 and 120kHz.


Please note that the above waveforms are simulated using a perfect square wave input, with rise and fall times of zero seconds. The output is measured between Q2 emitter and ground with a 200 ohm resistive load. In the real world, the cable to the remote photo emitter LEDs will contain both capacitance and inductance. This will increase both rise and fall times of the output signal. As with the Mark 1 circuit I recommend using speaker wire or bell wire to be used to cable the remote photo emitters.

My Prototype

Note that the veroboard layout below only includes the componets from the left of the schematic to C4, I had Q1 and Q2 on breadboard during this testing phase.

Setup and Testing:
There is little to adjust in this circuit. First I suggest disconnecting the wiring to the emitters IR2 and IR3. Switch on and D2 should be off. Aim a remote in the direction of IR1 and press any button D2 should light and be seen to flash when a button is held on the handset and go off when unpressed. If all is well reconnect the wiring to emitters IR2 and IR3. Without lenses, the light is quite directional and so you will need to aim it carefully at the remote equipment you are controlling. A digital camera, or camcorder can "see" into the Infra red range. This is useful to prove that IR2 and IR3 are producing output.

Veroboard Layout:
Below is a picture of my veroboard layout for the Mk 5 IR extender using Ron Js excellent veroboard images. Special thanks to Derek Smith for checking the veroboard layout and pointing out one small error (which is corrected now).


Special Note:I have omitted Diode D1 in my prototype and also the veroboard layout above, and the two images below. Click the links below to view the actual veroboard layouts. The veroboard drawing above shows the component site, the yellow circles represent the breaks on the bottom (track side).

Component side (106k)
Track side (97k)Note that this is reversed from component side.
For more help on vero layouts see this Practical Page.

Fault Finding:
If your circuit does not work, first check that your circuit is receiving power. Next compare the voltages to my prototype below. These checks are all made with a digital multimeter with a supply voltage of 12V DC. All checks are made with respect to ground (i.e. the back or negative meter probe is always connected to the negative or 0V power rail).

With no input signal:

IC1   Pin6       1.15V
IC2   Pin6       0V
Q1    base       0.8V
Q1    collector  0.13V
Q2    emitter    0V


With a strong input signal (handset same room less than 2meters away):

IC1   Pin6       1.15V
IC2   Pin6       0.15V
Q1    base       0.65V
Q1    collector  3.16V
Q2    emitter    2.79V



A good tip from Derek Smith (UK, who had problems with poor noise immunity in this circuit. Derek cured his fault by replacing the SFH2030 photo diode, the new SFH2030 provided much better noise immunity. So, if your voltage levels are similar to my prototype above then try replacing IR1.

If you still have problems with noise immunity check the supply voltage. Special thanks to Roch who found out that his 12V power supply was actually running at 16V. After reducing the voltage to 9V the problems disappeared for him. My original circuit ran happily from a 12V regulated supply.

Compatible Handsets:
If you build the mark 5 circuit please let me know the make and model of your remote control. I will add it to the list of compatible handsets below:-

Aiwa RC-ZVR01
Echostar URC-39756
Kameleon One for all remote (URC-8060) Maplins 6 way Audio/Video Switcher Hub order code L63AB
One for all remote
Panasonic EUR511200
Panasonic DVD player model no N2OAHC000012
Philips RC6512
Pioneer AXD7323
Pioneer VXX2801
Pioneer DVD remote
RCA systemlink 8 A-V
Saisho VR3300X
Sanyo vhs remote
Sony RM1- V141A VTR/TV
Sony RM-533
Sony RM-831
Std Sky digi box handset
Technics EUR64713
Xbox Remote

This is a compact three transistor, regenerative receiver with fixed feedback. It is similar in principle to the ZN414 radio IC which is now no longer available. The design is simple and sensitivity and selectivity of the receiver are good.

AM Receiver Schematic

Notes:
All general purpose transistors should work in this circuit, I used three BC109C transistors in my prototype.The tuned circuit is designed for medium wave. I used a ferrite rod and tuning capacitor from an old radio which tuned from approximately 550 - 1600kHz. Q1 and Q2 form a compund transistor pair featuring high gain and very high input impedance. This is necessary so as not to unduly load the tank circuit.

The 120k resistor provides regenerative feedback,between Q2 output and the tank circuit input and its value affects the overall performance of the whole circuit. Too much feedback and the circuit will become unstable producing a "howling sound". Insufficient feedback and the receiver becomes "deaf". If the circuit oscillates,then R1s value may be decreased; try 68k. If there is a lack of sensitivity, then try increasing R1 to around 150k. R1 could also be replaced by a fixed resisor say 33k and a preset resistor of 100k. This will give adjustment of sensitivity and selectivity of the receiver.

Transistor Q3 has a dual purpose; it performs demodulation of the RF carrier whilst at the same time, amplifying the audio signal. Audio level varies on the strength of the received station but I had typically 10-40 mV. This will directly drive high impedance headphones or can be fed into a suitable amplifier.

Construction:
All connections should be short, a veroboard or tagstrip layout are suitable. The tuning capacitor has fixed and moving plates. The moving plates should be connected to the "cold" end of the tank circuit, this is the base of Q1, and the fixed plates to the "hot end" of the coil, the juction of R1 and C1. If connections on the capacitor are reversed, then moving your hand near the capacitor will cause unwanted stability and oscillation.

Finally here are some voltagee checks from my breadboard prototype.This should help in determining a working circuit:

All measurements made with a fresh 9volt battery and three BC109C transistors with respect to the battery negative terminal.

Q1 (b) 1.31V
Q2 (b) 0.71V
Q2 (c) 1.34V
Q3 (b) 0.62V
Q3 (c) 3.87V

Notes:
To start in 24 seconds; 24s LOAD SW and Reset SW should be push simultaneously. If not, the count will start in 99. Pulse input can be connected to 555 astable multivibrator but must be calibrated for real time clock. The PAUSE SW must have a Switch Debouncer so that the counter will count normal when counting is paused and then turn-on.
When the count reach 00, the NOR gate will have an output of logic1 that will turn on the two transistor. The buzzer will rung and light will turn on. The two transistors are continuously turn-on not until LOAD SW and Reset SW is push. All have a +5v power supply.

Author: Milardo de Guzman
E-mail: milardo_dg@yahoo.com
Source: http://www.zen22142.zen.co.uk